An inverter has been known as a power conversion device configured to convert a direct-current voltage into an alternating-current voltage. Particularly, in recent years, as compared to a two-level inverter configured to obtain an alternating-current output by generating PWM pulse voltages of ±Ed having a zero point as a center from a direct-current voltage Ed, a three-level inverter has attracted more attention which is configured to obtain an alternating-current output by generating two types of PWM pulse voltages of ±Ed and ±(Ed/2) having a zero point as a center.
FIG. 5 depicts a schematic configuration of a so-called T-NPC (T-type Neutral Point Clamped) three-level inverter 1. The three-level inverter 1 is configured to switch a direct-current voltage Ed, which is applied between a direct-current high potential terminal P and a direct-current low potential terminal N, by a voltage clamped with a direct-current intermediate voltage (Ed/2), thereby generating the PWM pulse voltages of the two-step voltages in an alternating-current output terminal AC. The PWM pulse voltages are wave-filtered (filtering) by an LC filter (not shown), for example, so that an alternating-current voltage is generated. A waveform of the alternating-current voltage generated by the three-level inverter 1 is similar to a sinusoidal waveform smoother than a waveform of an alternating-current voltage that is to be generated by the two-level inverter.
Specifically, the three-level inverter 1 includes a first semiconductor switching element T1 provided between the direct-current high potential terminal P and the alternating-current output terminal AC and configured to be turned on/off during a positive voltage output mode, and a first reflux diode D1 connected reverse-parallelly with the first semiconductor switching element T1. Also, the three-level inverter 1 includes a second semiconductor switching element T2 provided between a direct-current low potential terminal N, which is paired with the direct-current high potential terminal P, and the alternating-current output terminal AC and configured to be turned on/off during a negative voltage output mode, and a second reflux diode D2 connected reverse-parallelly with the second semiconductor switching element T2. For reference, the first and second semiconductor switching elements T1, T2 are respectively configured by a high breakdown voltage IGBT (Insulated Gate Bipolar Transistor), for example. Also, the first and second reflux diodes D1, D2 are respectively configured by a bipolar diode based on Si.
Also, the three-level inverter 1 includes a bidirectional switch circuit BSW provided between a direct-current intermediate potential terminal M and the alternating-current output terminal AC. The bidirectional switch circuit BSW is configured by a so-called RB-IGBT (Reverse Blocking IGBT) in which third and fourth semiconductor switching elements T3, T4 each of which is configured by an IGBT having a reverse breakdown voltage are connected reverse-parallelly with each other. Also, the direct-current intermediate potential terminal M is applied with a direct-current intermediate voltage (Ed/2), which is obtained by dividing the direct-current voltage Ed by capacitors C1, C2 provided in series between the direct-current high potential terminal P and the direct-current low potential terminal N.
The bidirectional switch circuit BSW functions to change voltages, which are respectively applied to the first and second semiconductor switching elements T1, T2 by selectively applying the direct-current intermediate voltage (Ed/2) given to the direct-current intermediate potential terminal M to the alternating-current output terminal AC, by two steps as the voltage Ed or the voltage (Ed/2).
The basic operations of the three-level inverter 1 configured as described above are described in detail in JP-A-2012-130224, for example. For reference, a power conversion device configured to obtain a three-phase alternating-current output includes three sets of the three-level inverters 1 provided in parallel, and is configured to drive the three-level inverters 1 with a phase difference of 120°, thereby generating a three-phase alternating-current voltage consisting of U phase/V phase/W phase.
JP-A-2014-57520 suggests using, as the first and second reflux diodes D1, D2, a unipolar diode based on silicon carbide (SiC), which is a wide bandgap semiconductor, instead of the general bipolar diode of the related art based on silicon (Si). As disclosed in JP-A-2014-57520, when the unipolar diodes (SiC diodes) are used as the first and second reflux diodes D1, D2, it is possible to suppress a switching loss, i.e., a so-called recovery loss associated with reverse recovery operations of the first and second semiconductor switching elements T1, T2.
In the three-level inverter configured as described above, when the second semiconductor switching element T2 is turned off in a state where the direct-current voltage Ed is applied, for example, a reverse recovery current may flow for minor time of 1 μs or shorter via the first reflux diode D1, which is originally in an off state, in association with the reverse recovery operation of the second semiconductor switching element T2. This phenomenon similarly occurs when the first semiconductor switching element T1 is turned off in the state where the direct-current voltage Ed is applied. In this case, a reverse recovery current flows for minor time of 1 μs or shorter via the second reflux diode D2, which is originally in an off state, in association with the reverse recovery operation of the first semiconductor switching element T1.
For reference, the reverse recovery current flowing via the first reflux diode D1 is a factor of causing a high surge voltage Vcep in the second semiconductor switching element T2. When a circuit inductance of a current pathway is denoted as L and a turn-off current is denoted as [di/dt], the surge voltage Vcep is expressed as follows.Vcep=(Ed/2)+L×|di/dt|
Also, when the surge voltage Vcep is higher than the direct-current voltage Ed, the reverse recovery current after turn-off of the second semiconductor switching element T2, which originally flows toward the direct-current intermediate potential terminal M via the bidirectional switch circuit BSW, is likely to flow toward the direct-current high potential terminal P via the first reflux diode D1 configured by the unipolar diode. That is, since a circuit impedance (inductance component) between the direct-current intermediate potential terminal M and the alternating-current output terminal AC is greater than a circuit impedance (inductance component) between the direct-current high potential terminal P and the alternating-current output terminal AC, the turn-off current is likely to flow through the first reflux diode D1.
Due to the current, the steep voltage change is caused in the first reflux diode D1, so that the first reflux diode D1 may break down. Furthermore, due to the steep voltage change caused in the first reflux diode D1, a frequency component thereof is applied to the first semiconductor switching element via a parasitic capacitance Cres of the first semiconductor switching element T1, so that a steep change in gate voltage may be caused. The steep change in gate voltage is a factor of gate breakdown of the first semiconductor switching element T1. Also, the current flowing via the first reflux diode D1 is an occurrence factor of loss in the inverter.
Similarly, the turn-off current of the first semiconductor switching element T1, which flows toward the direct-current intermediate potential terminal M via the bidirectional switch circuit BSW, is likely to flow toward the direct-current low potential terminal N via the second reflux diode D2 configured by the unipolar diode. Therefore, upon the turn-off of the first semiconductor switching element T1, the similar problems to upon the turn-off of the second semiconductor switching element T2 are caused.